Modulation Method and Apparatus

ABSTRACT

A modulation circuit for use in a radiofrequency transmitter includes a local oscillator circuit configured to generate one or more local oscillator signals at a desired frequency and with a duty cycle at or about twenty-five percent, and a modulator configured to generate one or more modulated signals responsive to the one or more local oscillator signals and one or more baseband information signals. In at least one embodiment, the modulation circuit includes a modulator comprising a combined mixing and transconductance circuit that includes a transistor circuit for each baseband information signal serving as a modulation input to the modulator. Each transistor circuit comprises a first transistor driven by the baseband information signal and coupling a modulator output node to a corresponding transconductance element, and a second transistor driven by one of the one or more local oscillator signals and coupling the corresponding transconductance element to a signal ground node.

RELATED APPLICATIONS

This application is a divisional of patent application Ser. No.13/015,157, filed Jan. 27, 2011, which is a divisional of patentapplication Ser. No. 11/292,488, filed Dec. 2, 2005.

BACKGROUND OF THE INVENTION

The present invention generally relates to signal modulation, such asfor radiofrequency signal generation, and particularly relates tomodulation waveforms and corresponding modulation circuits.

Transmitters used in communication devices and systems commonly employmodulation circuits to up-convert baseband information signals to adesired (carrier) frequency. More particularly, such devices and systemsimpose transmit information on carrier signals, which usually aregenerated at desired or assigned transmit channel frequencies, bymodulating carrier signal phase, frequency, amplitude, or somecombination thereof, according to one or more baseband informationsignals representing the desired transmit information.

Quadrature modulation, also referred to as “IQ” modulation, uses twocarrier signals, an in-phase carrier and quadrature carrier that isoffset from the in-phase carrier by 90 degrees. The two carriersgenerally are modulated by corresponding in-phase and quadraturebaseband information signals and then combined for amplification andtransmission over a communication channel. IQ modulation findswidespread use in a variety of wireless communication systems, such asin cellular communication networks based on Wideband Code DivisionMultiple Access (WCDMA) or cdma2000 standards.

SUMMARY OF THE INVENTION

According to the methods and apparatus taught herein, one embodiment ofa modulation circuit comprises a local oscillator circuit configured togenerate one or more local oscillator signals at a desired frequency andwith a duty cycle at or about twenty-five percent, and a modulatorconfigured to generate one or more modulated signals responsive to theone or more local oscillator signals and one or more basebandinformation signals. The modulation circuit may be used, for example, ina wireless communication device, such as a cellular radiotelephone.

In one embodiment, the modulator comprises a transconductance stagecircuit configured to generate one or more current signals responsive tothe one or more baseband information signals, and a mixer stage circuitconfigured to generate one or more mixer output signals responsive tothe one or more current signals and the one or more local oscillatorsignals. In another embodiment, the mixer stage circuit and thetransconductance stage circuit are effectively combined. The resultingmodulator configuration offers a number of advantages, which, by way ofnon-limiting example, include reduced drive amplitude requirements andimproved output voltage swing.

One embodiment of the above-described combined mixing andtransconductance circuit includes a transistor circuit for each basebandinformation signal serving as a modulation input to the modulator. Eachsuch transistor circuit comprises a first transistor driven by thebaseband information signal and coupling a modulator output node to acorresponding transconductance element, and a second transistor drivenby one of the one or more local oscillator signals and coupling thecorresponding transconductance element to a signal ground or referencenode.

With these modulator variations in mind, one embodiment of a method ofimproving modulator operation is based on generating local oscillatorsignals at or about a twenty-five percent duty cycle. One such methodcomprises generating one or more local oscillator signals at the desiredfrequency and with a duty cycle of at or about twenty-five percent, anddriving corresponding local oscillator signal inputs of a modulator withthe one or more local oscillator signals. The modulator may employseparate transconductance and mixing stage circuits, or may employ acombined mixing and transconductance stage circuit.

Of course, the present invention is not limited to the above featuresand advantages. Those skilled in the art will recognize additionalfeatures and advantages upon reading the following detailed description,and upon viewing the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of one embodiment of a modulation circuit.

FIG. 2 is a schematic diagram of one embodiment of the modulatorillustrated in the modulation circuit of FIG. 1.

FIG. 3 is a group of waveforms illustrating the local oscillatorsignals, the baseband information signals and the modulated outputsignal of FIG. 2, for example,

FIG. 4 is a waveform graph illustrating one embodiment of generatinglocal oscillator signals having a twenty-five percent duty cycle.

FIG. 5 is a graph relating local oscillator signal duty cycle tosignal-to-noise-ratio.

FIG. 6 is a schematic diagram of another embodiment of the modulatorillustrated in the modulation circuit of FIG. 1.

FIG. 7 is a group of waveforms illustrating the local oscillatorsignals, the baseband information signals and the modulated outputsignal of FIG. 6, for example.

FIG. 8 is a schematic diagram of another embodiment of the modulatorillustrated in the modulation circuit of FIG. 1.

FIG. 9 is a block diagram of one embodiment of a wireless communicationdevice that includes an embodiment of the modulation circuit of FIG. 1.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 illustrates one embodiment of a modulation circuit 10, such asmight be implemented in a wireless communication device or system. Inthe illustrated embodiment, the modulation circuit 10 includes or isassociated with a local oscillator circuit 12 that is configured togenerate one or more local oscillator signals at a desired frequency andwith a duty cycle at or about twenty-five percent (as compared to theconventional fifty percent duty cycle used for such signals). Themodulation circuit 10 further includes a modulator 14, which isconfigured to generate one or more modulated signals responsive to theone or more local oscillator signals and one or more basebandinformation signals. In the diagram, the modulated signals, i.e., themodulated carrier signal(s) output by the modulator 14, are labeled as“modulator output signal(s),” and may comprise a single-ended mixeroutput signal or a differential pair of mixer output signals. In anycase, the modulator output signal is input to a power amplifier (PA)circuit 16 that is configured to amplify it for transmission.

In looking at the modulator 14 in more detail, one sees that theillustrated modulator embodiment comprises a transconductance stagecircuit 18 that is configured to generate one or more (differential)current signals responsive to the one or more baseband informationsignals, and further comprises a mixer stage circuit 20 that isconfigured to generate one or more mixer output signals responsive tothe one or more current signals and the one or more local oscillatorsignals. Note that the mixer output signals are generated by the mixerstage circuit 20 via a mixer load circuit 22, embodiments of which aredetailed later herein.

Further, note that the one or more baseband information signals aredepicted as differential pairs of in-phase and quadrature signals,labeled VIp/VIn to denoted the positive and negative or complementarysignal components. Likewise, the one or more local oscillator signalsare depicted as differential pairs of in-phase and quadrature localoscillator signals, labeled LOIp/LOIn, for the in-phase component, andlabeled LOQp/LOQn, for the quadrature component.

FIG. 2 provides a schematic illustration for one embodiment of themodulator 14 shown in FIG. 1. In particular, the illustrated circuitarrangement is adapted for the differential pairs of in-phase andquadrature baseband and local oscillator signals described above, andthus comprises two double-balanced mixers, wherein the mixer outputcurrents (Ic1 and Ic2) are added in the load.

In more detail, one sees that the transconductance stage circuit 18comprises a transistor M1 driven by one component of the differentialbaseband information signal pair VIp/VIn. The transistor M1, which isassociated with a corresponding transconductance element (shown as R1),converts the input baseband information signal into a current-modesignal coupled to the M2/M3 transistor pair of the mixer stage circuit20. The M2/M3 transistor pair is driven by the differential localoscillator signal pair LOIp/LOIn. The other component of the VIp/VIndifferential pair drives a transistor M4, which similarly is associatedwith a corresponding transconductance element (R2), and the M5/M6transistor pair in the mixer stage circuit 20.

With this arrangement, the differential current modulation generatedfrom the VIp/VIn baseband information signals are up-converted by theM2/M3 and M5/M6 transistor pairs of the mixer stage circuit 20 to thefrequency of the local oscillator signal LOIp/LOIn. A similararrangement and operation applies for quadrature baseband informationsignal VQp/VQn and the quadrature local oscillator signal LOQp/LOQn, viathe transistors M7 and M10, and their corresponding transconductanceelements R3 and R4, and their corresponding mixer stage transistor pairsM8/M9 and M11/M12.

Such operation produces a differential current Ic1 and Ic2 representinga carrier frequency signal having modulations corresponding to thebaseband information signals VIp/VIn and VQp/VQn. The differentialcurrent drives the mixer load circuit 22, which, in the illustratedembodiment, comprises the inductors L1 and L2, a capacitor C1, and aresistor R5. By operation of the mixer load circuit 22, the differentialcurrent Ic1 and Ic2 produce a voltage-mode, modulated carrier signal atthe input of the power amplifier 16, which amplifies it fortransmission. Note, too, that the mixer load circuit 22 generally isresonantly tuned to suppress harmonics of the local oscillatorfrequency, f_(LO).

Advantageously, the local oscillator circuit 12 is configured to providethe local oscillator signals, LOIp/LOIn and LOQp/LOQn, as switchingwaveforms at the desired frequency and with a duty cycle at or abouttwenty-five percent. FIG. 3 illustrates one embodiment of localoscillator signal waveforms, and further depicts corresponding basebandinformation signal and power amplifier output waveforms, for a givenconfiguration of the corresponding modulator circuit 14 and poweramplifier circuit 16. Note that it should be understood that theillustrated scale in FIG. 3 is representative rather than limiting.

With that in mind, the bottom-most waveform of FIG. 3 represents theLOIp component of the differential waveform pair, LOIp/LOIn. Itsdepicted amplitude is 2 Volts peak-to-peak, but it should be understoodthat the driving amplitude and the relative offset voltages used for theLOI (and LOQ) signals will be a matter of design, and depends on anumber of considerations, such as the configuration of the modulator 14,the transistor threshold voltages involved, the power supply and biasingvoltages used, the desired operating temperature range, and so on.

In any case, proceeding logically upward in the waveform diagram, thenext waveform depicted represents the differential waveform (LOIp−LOIn),and one sees that the twenty-five percent duty cycle generation of LOIpand LOIn produce a differential waveform having a characteristic steppedsquare wave. For the 2 V pk-pk generation of LOIp and LOIn, thedifferential component (LOIp−LOIn) will have an effective amplitude of−2 V to 2 V. It should be understood that a similar set of waveformpatterns apply to the quadrature baseband signal LOQp/LOQn.

Continuing with the diagram, the next two waveforms depicted representthe differential baseband information signals (VIp−VIn) and (VQp−VQn).Finally, the topmost waveform represents the correspondingly modulatedoutput signal, VRF_OUT, as generated by the power amplifier circuit 16.

With the above details in mind, one may appreciate at least some of theadvantages of operating the modulator 14 at or about a twenty-fivepercent duty cycle, as compared to the conventional operation at orabout fifty percent duty cycles. Operating with a twenty-five percentduty cycle, such as represented for one or more embodiments by thetwenty-five percent duty cycles of the local oscillator waveformsdepicted in FIG. 4, reduces the conversion gain of the modulationcircuit 10 by three 3 dB, but doing so also halves the currentconsumption of the modulator 14 and halves the noise of that circuit.Halving the current consumption reduces the output power by 3 dB, butthe corresponding reduction in noise means that the signal-to-noiseratio (SNR) of the mixer output signal remains the same. Alternatively,one could operate the modulator 14 at a twenty-five percent duty cyclebut with a current consumption comparable to operation at a fiftypercent duty cycle, and thereby improve the output power and SNR by 3dB.

Understanding the above details begins with a generalized Fourier seriesrepresentation of the local oscillator signal waveforms depicted in FIG.4, for example, which is given as

$\begin{matrix}{{s(t)} = {\frac{\tau}{T} + {2\frac{\tau}{T}{\sum\limits_{n = 1}^{\infty}{\frac{\sin \left( {n\; \omega_{LO}\frac{\tau}{2}} \right)}{n\; \omega_{LO}\frac{\tau}{2}}{\cos \left( {n\; \omega_{LO}t} \right)}}}}}} & \left( {{Eq}.\mspace{14mu} 1} \right)\end{matrix}$

where

$T = {\frac{2\pi}{\omega_{LO}}.}$

Expressing the duty cycle term as

$\eta = \frac{\tau}{T}$

allows, for example, the LOIp(t) waveform of FIG. 4 to be expressedgenerally as

$\begin{matrix}{{s_{p}(t)} = {\eta + {2{\sum\limits_{n = 1}^{\infty}{\frac{\sin \left( {n\; {\pi\eta}} \right)}{n\; \pi}{\cos \left( {n\; \omega_{LO}t} \right)}}}}}} & \left( {{Eq}.\mspace{14mu} 2} \right)\end{matrix}$

Thus, LOIn(t), being 180 degrees out-of-phase with LOIp(t), can beexpressed generally as

$\begin{matrix}{{s_{n}(t)} = {\eta + {2{\sum\limits_{n = 1}^{\infty}{\frac{\sin \left( {n\; {\pi\eta}} \right)}{n\; \pi}{\cos \left( {{n\; \omega_{LO}t} + {n\; \pi}} \right)}}}}}} & \left( {{Eq}.\mspace{14mu} 3} \right)\end{matrix}$

The differential signal will then be

$\begin{matrix}{{s_{diff}(t)} = {{{s_{p}(t)} - {s_{n}(t)}} = {\frac{4}{\pi}{\sum\limits_{n = 0}^{\infty}{\frac{\sin \left( {\left( {{2n} + 1} \right)\; {\pi\eta}} \right)}{{2n} + 1}{\cos \left( {\left( {{2n}\; + 1} \right)\omega_{LO}t} \right)}}}}}} & \left( {{Eq}.\mspace{14mu} 4} \right)\end{matrix}$

For a fifty percent duty cycle, i.e., η=½, (Eq. 4) yields

$\begin{matrix}{{s_{diff}(t)} = {\frac{4}{\pi}{\sum\limits_{n = 0}^{\infty}{\frac{\sin \left( {\left( {{2n} + 1} \right){\pi/2}} \right)}{{2n} + 1}{\cos \left( {\left( {{2n}\; + 1} \right)\omega_{LO}t} \right)}}}}} & \left( {{Eq}.\mspace{14mu} 5} \right)\end{matrix}$

For a twenty-five percent duty cycle, i.e., η=¼, (Eq. 4) yields

$\begin{matrix}{{s_{diff}(t)} = {\frac{4}{\pi}{\sum\limits_{n = 0}^{\infty}{\frac{\sin \left( {\left( {{2n} + 1} \right)\; {\pi/4}} \right)}{{2n} + 1}{\cos \left( {\left( {{2n}\; + 1} \right)\omega_{LO}t} \right)}}}}} & \left( {{Eq}.\mspace{14mu} 6} \right)\end{matrix}$

With the fifty percent duty cycle term of (Eq. 5) in mind, the currentconversion of the modulator 14 (from the differential outputs of thetransconductance stage 18 to the load currents Ic1 and Ic2) can bederived as

$\begin{matrix}{I_{out} = {\frac{I_{c\; 1} - I_{c\; 2}}{2} = {I_{{in},{diff}}\frac{2}{\pi}{\sum\limits_{n = 0}^{\infty}{\frac{\sin \left( {\left( {{2n} + 1} \right)\; {\pi/2}} \right)}{{2n} + 1}{\cos \left( {\left( {{2n}\; + 1} \right)\omega_{LO}t} \right)}}}}}} & \left( {{Eq}.\mspace{14mu} 7} \right)\end{matrix}$

Where, I_(in,diff) is the differential current input to the mixer stage20, which is the same as the differential output current of thetransconductance stage 18.

Now, for the twenty-five percent duty cycle term of (Eq. 6) in mind, theoutput current is

$\begin{matrix}{I_{out} = {\frac{I_{c\; 1} - I_{c\; 2}}{2} = {I_{{in},{diff}}\frac{2}{\pi}{\sum\limits_{n = 0}^{\infty}{\frac{\sin \left( {\left( {{2n} + 1} \right)\; {\pi/4}} \right)}{{2n} + 1}{\cos \left( {\left( {{2n}\; + 1} \right)\omega_{LO}t} \right)}}}}}} & \left( {{Eq}.\mspace{14mu} 8} \right)\end{matrix}$

Assuming that the mixer load circuit 22 is resonant at the localoscillator frequency, f_(LO), the amplitude of the fundamental tone(n=0) in the load current is

$I_{{in},{diff}}\frac{\sqrt{2}}{\pi}$

for twenty-five percent duty cycle operation. In comparison, theamplitude of the fundamental tone (n=0) in the load current of (Eq. 7),which, assuming the conventional use of fifty percent duty cycles in thelocal oscillator signals, is

$I_{{in},{diff}}{\frac{2}{\pi}.}$

With these expressions in mind, one sees that the modulator 14 can beoperated at a twenty-five percent duty cycle with a 3 dB reduction inpower but no degradation in SNR because of the corresponding reductionin current and noise

FIG. 5 plots

$\frac{\sin^{2}\left( {\pi \; \eta} \right)}{\eta}$

as a function of duty cycle η and further illustrates one or more of theadvantages of operating the modulator 14 at a twenty-five percent dutycycle. More particularly, from the fundamental tone of (Eq. 4), one seesthat the output power of the modulator is proportional to sin² (πη),wherein, in turn,

$\frac{\sin^{2}\left( {\pi \; \eta} \right)}{\eta}$

is proportional to SNR. Note, too, that the noise power is directlydependent on the duty cycle η. Further, one sees that the plotted ratiois maximized at approximately 0.37. However, a duty cycle ratio of 0.37is impractical, or at least decidedly more complex to generate than aduty cycle of 0.25. Indeed, a duty cycle of 0.25 can be reliably andcleanly generated by the local oscillator circuit 12 from an input clocksignal of the desired frequency, or at some multiple thereof, usingflip-flops and digital delay gates, for example. Further, one may use aJohnson-counter, a quadrature Voltage Controlled Oscillator (VCO), or adivide-by-2 circuit in combination with supporting logic.

Regardless of the generation details embodied in the local oscillatorcircuit 12, one sees that the plotted ratio has the same value at a dutycycle of 0.25 as it does at the conventionally used duty cycle of 0.5.This constancy in SNR as compared to conventional operation at fiftypercent duty cycles arises because the modulator's transconductancecurrent sources are on for half as much time and the total noise powertransferred to the mixer circuit load 22 is therefore halved.

The same or similar twenty-five percent duty cycle signals may beapplied to the embodiment of modulator 14 depicted in FIG. 6, and itshould be understood that this embodiment of the modulator 14 can besubstituted into the modulation circuit 10 of FIG. 1. Of course, itshould be further understood that the modulator 14 depicted in FIG. 6has advantages over the embodiment of modulator 14 as depicted in FIG. 2that are not dependent on providing it with local oscillator signalshaving a twenty-five percent duty cycle. However, it does offerexcellent performance when used in that context.

In more detail, the modulator 14 depicted in FIG. 6 includes a combinedmixer and transconductance stage circuit 28 that provides, among otherthings, improved output voltage swing capability of the modulator 14.Increased output voltage swing increases the output power of themodulator 14 for the same input power—i.e., improves its efficiency.

One notable feature of the combined mixer and transconductance stage 28depicted in FIG. 6 is that the switching transistors driven by the localoscillator signals are moved to the bottom of the transconductancetransistors driven by the baseband information signals. Moreparticularly, the combined mixer and transconductance stage 28 includesa transistor circuit for each baseband information signal serving as amodulation input to the modulator 14. Each transistor circuit comprisesa first transistor driven by the baseband information signal andcoupling a modulator output node to a corresponding transconductanceelement, and a second transistor driven by one of the one or more localoscillator signals and coupling the corresponding transconductanceelement to a signal ground node.

This arrangement is plainly seen, for example, wherein the transistorM14 represents the first transistor of one of the transistor circuits,the resistor R6 represents the corresponding transconductance element,and the transistor M15 represents the second transistor of the sametransistor circuit. Further, one sees that transistor M14 is driven byone of the baseband information signals—depicted as the positivecomponent of the differential baseband signal pair VIp and VIn—and thattransistor M14 couples one of the modulator output nodes 30 and 32 toone end of the transconductance element R6. The other end of R6 iscoupled to a signal ground or reference node 34 through transistor M15,which is driven by one of the local oscillator signals—depicted as thepositive component of the differential local oscillator signal pair LOIpand LOIn. Similar operation is provided by the transconductancetransistors M16-M28 (even), and their corresponding transconductanceelements R7-R13 and switching transistors M15-M29 (odd). Note, too, thatthe switching transistors M15-M29 (odd) can be placed on the drains orgates of the transconductance transistors M14-M28 (even).

In any case, for differential signal configurations, the combined mixingand transconductance circuit 28 generally includes a pair of theillustrated transistor circuits for each differential signal in thedifferential signal pair. While this configuration effectively doublesthe number of current sources as compared to the transconductance stagecircuit 18 illustrated for the modulator embodiment of FIG. 2, therelative current levels are halved and SNR is thus maintained. Further,as noted above, the illustrated circuit arrangement provides improvedoutput voltage swing. More particularly, one sees that in, FIG. 2, theswitching transistors—the transistors driven by the local oscillatorsignals—appear between the mixer load and the transconductancetransistors—the transistors driven by the baseband information signals.That circuit arrangement means that the switching transistors, i.e.,transistors M2, M3, M5, M6, M8, M9, M11, and M12 of FIG. 2, consume afew hundred millivolts of output voltage headroom. In contrast, theswitching transistors M15-M29 (odd) in FIG. 6 are included below thetransconductance elements (resistors R6-R13) and, thus, are effectivelysubsumed in those resistances.

FIG. 7 illustrates one embodiment of local oscillator and basebandsignal waveforms that may be used with the modulator 14 of FIG. 6. Aswith the waveforms in FIG. 3, FIG. 7 illustrates in bottom-up order, theLOIp signal, the LOIp−LOIn differential signal, the VIp−Vin differentialsignal, the VQp−VQn differential signal, and the VRF_OUT signal. Ofcourse, those skilled in the art will appreciate that the illustratedsignal levels are provided only by way of non-limiting example.

However, it should be noted that positioning the switching transistorsM15-M29 (odd) at the bottom of the transconductance transistors M14-M28(even) offers, among other things, the advantage of using-reducedamplitudes for the local oscillator signals. To understand this, one mayrefer to the switching transistors of FIG. 2, which are shown in themixer stage circuit 20 as transistors M2/M3, M5/M6, M8/M9, and M11/M12.The drive voltages applied to the gates of those transistors must exceedthe gate-to-source threshold voltage by a sufficient margin to insurefull turn-on of the transistors. Because the gate-to-source voltage ofthose transistors is referenced to the voltage appearing on respectivesource nodes of the transconductance stage 18, the applied drivevoltages of LOIp/LOIn and LOQp/LOQn must be higher than the applieddrive voltages for the combined mixer and transconductance stage circuit28 of FIG. 6. That is, the gate-to-source voltage of the switchingtransistors M15-M29 (odd) in FIG. 6 is referenced to a common signalground node 34, meaning that these transistors can be fully turned onusing a comparatively lower amplitude for the local oscillator signals,independent of duty cycle.

With these and other advantages in mind, FIG. 8 illustrates anotherembodiment of the modulator 14, wherein a combined mixer andtransconductance stage circuit 40 is configured for singled-endedoperation. Here, a first transistor circuit comprises a first transistorM30, which couples a common modulator output node 36 to a correspondingtransconductance element R15. In turn, that transconductance element iscoupled to a common signal reference node 38 through a second transistorM31. The transistor M30 is driven by one component of the differentialbaseband information signal pair VIp/VIn, and the transistor M31 isdriven by one component of the differential local oscillator signal pairLOIp/LOIn. Transistors M32/M33, M34/M35, M36/M37, and theircorresponding transconductance elements R16, R17, and R18, providesimilar functionality for the remaining components of the differentialsignals VIp/VIn, VQp/VQn, LOIp/LOIn, and LOQp/LOQn.

The mixer load circuit 22 complements the single-ended configuration ofthe combined mixer and transconductance stage circuit 40 by providing asingle-ended connection between a supply voltage (VCC) and the commonmodulator output node 36. While it should be understood that the mixerload circuit 22 can be configured differently, the illustratedembodiment comprises a parallel RLC circuit that includes C3, L5, andR19.

It also should be understood that the modulation circuit 10 can be usedin a variety of applications involving the generation and transmissionof modulated signals, such as in the types of radiofrequencytransceivers used in wireless communication base stations and mobilestations. For example, FIG. 9 illustrates one embodiment of a wirelesscommunication device 50 that includes an embodiment of the modulationcircuit 10 described herein. Note, however, that various elements of themodulation circuit 10 may be distributed within the communication device50, rather than being wholly integrated together.

In the illustrated embodiment, the wireless communication device 50,which may be, for example, a cellular radiotelephone, PDA, wirelesspager, etc., comprises a transmit/receive antenna 52, a switch/duplexer54, a receiver 56, a transmitter 58, baseband processing circuits 60,system control circuits 62, input/output (interlace) circuits 64, anduser interface circuits 66. As will be understood, the user interfacecircuits 66 will vary according to the intended function of the device50, and thus may include a display screen, keypad, microphone, and aspeaker.

Regardless, the baseband processing circuits 60, which may comprise oneor more general- or special-purpose microprocessors and correspondingprogram instructions, may be configured to generate baseband informationsignals—e.g., quadrature signals—for input to the modulator 14 of thetransmitter 58. In turn, the local oscillator circuit 12, which mayoperate under command/control by the baseband processing circuits 60,can be configured to generate local oscillator signals for input to themodulator 14, which thus provides the power amplifier circuit 16 of thetransmitter 58 with one or more carrier signals for transmission thatare modulated in accordance with the baseband information signals.

With the above range of applications and embodiments in mind, it shouldbe understood that the present invention is not limited by the foregoingdescription, nor is it limited by the accompanying drawings. Instead,the present invention is limited only by the following claims, and theirlegal equivalents.

1. A modulation circuit for use in a radiofrequency transmitter, saidmodulation circuit comprising: a local oscillator circuit configured togenerate one or more local oscillator signals at a desired frequency andwith a duty cycle at or about twenty-five percent; and a modulatorconfigured to generate one or more modulated signals responsive to theone or more local oscillator signals and one or more basebandinformation signals.
 2. The modulation circuit of claim 1, wherein themodulator comprises a transconductance stage circuit configured togenerate one or more current signals responsive to the one or morebaseband information signals, and a mixer stage circuit configured togenerate one or more mixer output signals responsive to the one or morecurrent signals and the one or more local oscillator signals.
 3. Themodulation circuit of claim 1, wherein the modulator comprises acombined mixing and transconductance circuit that includes a transistorcircuit for each baseband information signal serving as a modulationinput to the modulator, each said transistor circuit comprising: a firsttransistor driven by the baseband information signal and coupling amodulator output node to a corresponding transconductance element; and asecond transistor driven by one of the one or more local oscillatorsignals and coupling the corresponding transconductance element to asignal ground node.
 4. The modulation circuit of claim 1, wherein thelocal oscillator circuit is configured to generate the one or more localoscillator signals as in-phase and quadrature signals.
 5. The modulationcircuit of claim 1, wherein the local oscillator circuit is configuredto generate the one or more local oscillator signals as transistor drivesignals for transistor-based switches in a mixing portion of themodulator.
 6. The modulation circuit of claim 1, wherein the modulatorcomprises one or more transconductance transistors driven by the one ormore baseband information signals and one or more switching transistorsdriven by the one or more local oscillator signals.
 7. A wirelesscommunication device including a modulation circuit comprising: a localoscillator circuit configured to generate one or more local oscillatorsignals at a desired frequency and with a duty cycle at or abouttwenty-five percent; and a modulator configured to generate one or moremodulated signals responsive to the one or more local oscillator signalsand one or more baseband information signals.
 8. The wirelesscommunication device of claim 7, wherein the modulator comprises acombined mixing and transconductance circuit that includes a transistorcircuit for each baseband information signal serving as a modulationinput to the modulator, each said transistor circuit comprising a firsttransistor driven by the baseband information signal and coupling amodulator output node to a corresponding transconductance element, and asecond transistor driven by one of the one or more local oscillatorsignals and coupling the corresponding transconductance element to asignal ground node.
 9. The wireless communication device of claim 8,wherein the baseband information signal comprises a differential signalpair, and wherein the combined mixing and transconductance circuitincludes a pair of the transistor circuits for each differential signalin the differential signal pair.